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1. (WO2018039674) WAKE UP RECEIVER USING MULTIPHASE PEAK DETECTOR AND DEMODULATOR
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WAKE UP RECEIVER USING MULTIPHASE PEAK DETECTOR AND DEMODULATOR

[0001] This relates generally to wake-up receivers, and more particularly to a wake-up receiver using a multiphase peak detector and demodulator.

BACKGROUND

[0002] A wake-up receiver is an auxiliary receiver that continuously monitors for communication requests, while allowing a main receiver to remain in sleep mode until communication is desired. Low latency wireless communication is the key to success for low power wake up receivers. Low latency can be achieved by consuming significant amounts of power in the front-end electronics, but this reduces battery longevity or limits the signal processing if the application operates with wireless charging. Conventional solutions for detecting a wake-up request have used a logarithmic amplifier to detect the signal at radio frequency (RF), pattern detection with a mixer followed by pseudo-differential stages with cascaded high-pass and low-pass frequency responses, and energy detection with respect to a single phase of the signal.

SUMMARY

[0003] In described examples, a wireless receiver includes a peak detector coupled to receive phases of an input signal and to provide a detector output. The phases are substantially in quadrature. Also, the wireless receiver includes a demodulator, which includes an analog comparator coupled to receive the detector output and to provide a comparator output.

BRIEF DESCRIPTION OF THE DRAWINGS

[0004] FIG. 1 depicts a circuit diagram of a quadrature phase peak detector according to an embodiment.

[0005] FIGS. 1A and IB depict the input signals and the output signals respectively of the circuit of FIG. 1 according to an embodiment.

[0006] FIG. 2 depicts a circuit diagram of a quadrature phase peak detector according to an embodiment.

[0007] FIGS. 2A and 2B depict the input signals and the output signal respectively of the circuit of FIG. 2 according to an embodiment.

[0008] FIG. 3 A depicts a circuit diagram of a current comparator used with a quadrature phase peak detector according to an embodiment.

[0009] FIG. 3B depicts the circuit of FIG. 3A modified to provide differential outputs according to an embodiment.

[0010] FIG. 3C depicts the differential outputs of the circuit of FIG. 3B according to an embodiment.

[0011] FIG. 3D depicts an alternative version of the circuit that provides differential outputs according to an embodiment.

[0012] FIG. 4 depicts the way in which example output signals from two successive stages of peak detectors can be used to provide a digital output according to an embodiment.

[0013] FIG. 5A depicts a block diagram of a low power wake-up receiver according to an embodiment.

[0014] FIG. 5B depicts a block diagram of a demodulator according to an embodiment.

[0015] FIG. 5C depicts a block diagram of a low power wake-up receiver according to an embodiment.

[0016] FIG. 6 depicts a circuit diagram of a passive asynchronous quadrature phase shifter according to an embodiment.

[0017] FIG. 7A depicts a block diagram of a low power wake-up receiver according to an embodiment.

[0018] FIG. 7B depicts a block diagram of a low power wake-up receiver according to an embodiment.

DETAILED DESCRIPTION OF EXAMPLE EMB ODEVIENT S

[0019] In the drawings, like references indicate similar elements. In this description, different references to "an" or "one" embodiment are not necessarily to the same embodiment, and such references may mean at least one. Further, when a particular feature, structure or characteristic is described in connection with an embodiment, such feature, structure or characteristic may be effected in connection with other embodiments, irrespective of whether explicitly described.

[0020] Example embodiments implement demodulators that operate using multiple phases of input signals, specifically, using quadrature phases of the signals. Conventional ASK demodulators utilize either a single phase or differential phase signals. Use of N phases reduces the detection time by a factor of N by decreasing the settling time of a peak detector. Due to the inherent low ripple resulting from the N phases, the filtering elements also require low area on chip (i.e., the cut-off frequency of these filters becomes N times higher compared to a single peak detector) Also, distributed demodulators receive signals after each gain/filter stage. With each stage providing gain to the desired signal and filtering to the undesired signal over the previous stage, an output from one stage can be used as a threshold value to the output from the following stages, providing a threshold value that is automatically dependent on the signal amplitude, so that a precise reference voltage for the threshold value is not necessary. These improvements can be incorporated into an overall architecture for a low power wake-up receiver.

[0021] Example embodiments include a system that uses quadrature copies of an input signal to allow faster detection of a signal and a demodulator that does not require a precise threshold to be applied to determine the digital value of the signal. Embodiments of the overall system convert an input signal to quadrature signals before providing the input signal to the demodulator. As described hereinbelow, in one embodiment, two or more mixers mix the incoming differential signals with quadrature frequencies in order to provide quadrature signals to filtering stages, ensuring that processing is largely performed in quadrature. In another embodiment, the mixers and filter stages use differential signals and then convert the differential signals to quadrature signals using a passive asynchronous quadrature phase shifter before providing the signals to demodulators.

[0022] In at least one example, a demodulator requires at least a comparator and a translator circuit that determines a digital value corresponding to the comparator output. A peak detector can be regarded as either a module that provides input to the demodulator or else an early stage of the demodulator, as these circuits can be closely linked with each other. FIG. 1 depicts a quadrature phase peak detector 100, which is part of the demodulator in at least one embodiment. In this drawing, MOS transistors 102, 104, 106, and 108 are coupled in parallel, with their respective drains coupled to VDD and their sources coupled to connector 114, which is also coupled to current sink IBIAS- Resistor R has a first terminal coupled to connector 114 and a second terminal coupled to capacitor C, which has its other terminal coupled to local ground. VOUT can be taken between resistor R and capacitor C. Although transistors 102, 104, 106, and 108 are implemented as NMOS transistors in this embodiment, they may also be implemented as PMOS transistors.

[0023] Each of transistors 102, 104, 106, 108 has its gate controlled by one of four quadrature signals, which are shown in FIG. 1A and which originate in the filtering stages of the receiver. These four signals include signal Φι, which is identified as 1+ (0°), signal Φ2, identified as I. (180°), signal Φ3, identified as Q+ (90°), and signal Φ4, identified as Q. (270°). The gate of transistor 102 is controlled by signal Φι, while the gates of transistors 104, 106 and 108 are controlled respectively by signals Φ2, Φ3, and Φ4. Although not specifically shown in this drawing, each of transistors 102, 104, 106, 108 is biased at close to its threshold voltage, i.e., gate-source voltage VGS is approximately equal to the threshold voltage VTH- In operation, connector 114 provides signal 110 and VOUT provides signal 112, both of which are shown in FIG. IB.

[0024] By controlling the gate of transistors 102, 104, 106, 108 with the quadrature signals and taking the output from the source, quadrature phase peak detector 100 provides high input impedance and low output impedance. Providing at least one low impedance point in the peak detector reduces a time constant associated with operation of the peak detector. The multiple phases of the input signal provide less ripple in the output signal. Consequently, in one embodiment, the signal settles in less than 500 microseconds with a robust scheme and good sensitivity and selectivity. Due to the low ripple or noise, a receiver using the described peak detector may require less filter area and use a smaller capacitor than previously required. The multiple phases do not require a high degree of accuracy in the spread between signals, i.e., if instead of 0° and 90°, the signals are 0° and 95°, the effect on the results are negligible, so no calibration is needed.

[0025] FIG. 2 depicts quadrature phase peak detector 200 according to an alternative embodiment. In this embodiment, MOS transistors 202, 204, 206, 208 are again coupled in parallel, although in this embodiment, the quadrature signals Φι, Φ2, Φ3, and Φ4 (shown again in FIG. 2A) are applied to the source terminals of transistors 202, 204, 206, 208 respectively. The drain terminals of these four transistors are each coupled to connector 214, which in turn is coupled to the upper rail through resistor R. Capacitor C has a first terminal coupled to connector 214 and a second terminal coupled to local ground. VOUT is taken from connector 214 and provides output signal 212, shown in FIG. 2B. As in the embodiment described hereinabove, each of transistors 102, 104, 106, 108 is biased at close to its threshold voltage (not specifically shown).

[0026] Quadrature phase peak detector 200 exhibits current mode input with low input

impedance and voltage mode output with high output impedance. As described hereinabove, providing at least one low impedance point in the analog detector reduces the time constant associated with operation of the peak detector. As in the examples described hereinabove, transistors 202, 204, 206, 208 are not limited to NMOS transistors, but may also be PMOS transistors. This embodiment shares the advantages of quadrature phase peak detector 100, i.e., low ripple, and settles in less than 500 microseconds, a latency that is related to the period of the signal applied to the peak detector. In both of these embodiments, the use of N phases artificially creates an apparent frequency of N*Fc, where Fc is the frequency of the input signal. Given the lower ripple, filtering components may be smaller or in some embodiments may be eliminated to save area. This technique can easily be extended into eight or more phases, as will be described hereinbelow. In such an embodiment, quadrature phase peak detectors 100, 200 will each include N transistors corresponding to the N phases.

[0027] FIG. 3A discloses a current comparator 300A according to an embodiment. Current comparator 300 A incorporates quadrature phase peak detector 100 containing transistors 102, 104, 106, 108, each receiving one of quadrature signals Φι, Φ2, Φ3, and Φ4 on a respective gate terminal, and also includes transistors 310, 312, 314, 316, 318, 320. The source of PMOS transistor 312 is coupled to the upper rail and the drain is coupled in series with NMOS transistors 316 and 320 to ground. PMOS transistor 310 has its source coupled to the upper rail and both its gate and drain coupled to drain connector 322 of peak detector 100 and NMOS transistors 314 and 318 are coupled in series between source connector 114 of peak detector 100 and the lower rail. The gates of PMOS transistors 310 and 312 are coupled together, with resistor R and one terminal of capacitor C coupled between the two transistors; a second terminal of capacitor C is coupled to the lower rail. NMOS transistors 314 and 316 have gates coupled to input VBIASI and NMOS transistors 318 and 320 have gates coupled to input VBIAS2- These input bias values define operating conditions for the associated transistors and provide a low voltage value that overcomes the threshold value of the respective transistors and allows more efficient operation. Together, transistors 310, 312, 314, 316, 318, and 320 form a current mirror such that PMOS transistor 312 mirrors the drain current from peak detector 100 and NMOS transistors 316, 320 mirror the source current from peak detector 100. Output signal VOUT reflects a difference between the two drain currents of transistors 312 and 316 that is proportional to the envelope of the input signal.

[0028] FIG. 3B discloses a current comparator 300B, which provides differential outputs according to an embodiment. The upper portion of the circuit for comparator 300B is the same as for comparator 300 A, except that PMOS transistor 312 has its source connected to ground, with VOUT+ being taken between the drain of transistor 312 and ground. Below peak detector 100, MOS transistors 314, 316 have their gates tied together and also tied to connector 114 to create a separate current mirror. VOUT- is taken between the drain of NMOS transistor 316 and the lower rail. FIG. 3C illustrates the output of comparator 300B, which are differential signals. FIG. 3D discloses an alternative version of the current comparator of FIG. 3B in which a current mirror is not used. In this embodiment, peak detector 100 is connected to the upper rail through a first resistor and is connected to the lower rail through both a second resistor and a capacitor connected in parallel. In this embodiment, VOUT+ is taken directly from drain connector 322 and VOUT- is taken directly from source connector 114.

[0029] FIG. 4 depicts a voltage mode comparator scheme that does not require a precise threshold to be set. When converting a signal from analog to digital, precise threshold values are generally generated on-chip from known references, such as bandgap etc., to provide a an average value to slice logic 1 and 0 levels. Setting the reference level equal to the average value leads to maximum noise margin in the 0 and 1 states. In an example system in which signal values can vary between zero and three volts, the threshold would generally be set at 1.5 volts. These architectures for a wake-up receiver use a distributed set of N analog demodulators in which each successive demodulator is coupled to receive the output from successive gain/filter modules. The output signal of each N-phase demodulator is proportional to the input signal amplitude, and all demodulators use the same input DC voltage, which is usually set at mid-rail. Accordingly, inputs to demodulator N and demodulator N+k, (where k>=l) are simply scaled versions of each other, and these two values can be put to a simple comparator, with the smaller value being used as a threshold value for the larger value, to provide rail to rail output voltage. In FIG. 4, the output signal from two successive demodulators is shown. Signal 402 is produced by a first demodulator N and signal 404 is produced by a successive demodulator N+k, whose signal has undergone further gain and filtering and consequently has a larger swing in the signal. By using signal 402 as a threshold for signal 404, generation of a precise on-chip threshold is not required. Instead, each demodulator provides an output that can be used as a threshold signal in a comparator associated with a successive demodulator output. The detector output of an ΝΛ

peak detector can also be compared to a linear combination of comparator outputs from a number of previous stages. In one embodiment, a linear combination of previous comparator outputs is determined using the formula, DO,N = VPD(N) - VpD(N-k) where DO,N denotes the arator output of an NTH stage, k is an integer less than N, and VPD,N denotes a detector output of an ΝΛ stage. A final output value is taken from the final demodulator stage and used to provide signal 406; preceding demodulators are used primarily to provide this threshold value to successive demodulators.

[0030] FIG. 5A shows a block diagram of a low power wake-up receiver 500A. Wake-up receiver 500A uses a multi-phase ASK demodulator that employs the described quadrature phase peak detector and also uses the output of a preceding demodulator as a threshold for a later demodulator. Signal processing is performed using synchronous quadrature phases and the mixers operate with quadrature switching waveforms that are precisely generated using frequency dividers inside the phase locked loop. Wake-up receiver 500 receives differential signals from antenna 512. These differential signals are received and amplified at amplifier 506 and passed to mixers 502A and 502B. Mixer 502A mixes the differential signals with 0° and 180° signals from a first local oscillator 520A and mixer 502B mixes the differential signals with 90° and 270° signals from a second local oscillator 520B to create two sets of differential signals that are in quadrature.

[0031] The differential signals from mixer 502A are passed successively to gain/filter stages 508A, 508B, 508C, and 508D while the quadrature signals from mixer 502B are passed successively to gain/filter stages 508E, 508F, 508G, and 508H. Each successive gain/filter stage 508 provides both filtering, e.g. to remove blocker signals, and additional gain to the signals passed there through. Gain/filter stages 508A and 508E are baseband gain/filter stages with precise gain; gain/filter stage 508A is followed by passive polyphase gain/filter stages 508B, 508C, 508D and gain/filter stage 508E is followed by stages 508F, 508G, 508H. The two baseband filter stages 508A, 508E are each used to suppress the carrier signal and each of polyphase gain/filter stages 508B, 508C, 508D 508F, 508G, 508H can be used to provide a null for a blocker signal. The entire architecture is fully differential up to the demodulator stage and provides high immunity to common mode noise.

[0032] Gain/filter stages 508A and 508E are coupled to provide quadrature inputs to demodulator 51 OA; gain/filter stages 508B and 508F are coupled to provide quadrature inputs to demodulator 510B; gain/filter stages 508C and 508G are coupled to provide quadrature inputs to demodulator 5 IOC; and gain/filter stages 508D and 508H are coupled to provide quadrature inputs to demodulator 510D. In the block diagram of this drawing, the peak detector is not specifically shown, but can be a part of the demodulator. This is illustrated in FIG. 5B, which provides a block diagram of demodulator 510 as shown in FIG. 5 A. This drawings shows that demodulator 510 includes peak detector 532, which receives quadrature input signals 516 and provides detector output signal(s) 540 to analog comparator 534. Analog comparator 534 in turn provides signal 542 to translator circuit 536 and translator circuit 536 determines whether the signal 542 should be considered a zero or a one and provides digital output signal 514. Because the analog detector in the example embodiment is operating with four phases, detection leads to four times faster detection compared to single phase operation. Also, the output of demodulator 510 is a stable square waveform compared to a threshold based scheme providing pulse outputs. This also adds to the benefit of the low latency scheme, which simplifies the digital demodulator back end. An output signal from each demodulator may be provided to a successive demodulator in the string of demodulators, i.e., demodulator 510A may provide output signal 514A to demodulator 510B; demodulator 510B may provide output signal 514B to demodulator 5 IOC; and demodulator 5 IOC may provide output signal 514C to demodulator 510D. In each of these instances, the output signal from one stage may be used as a threshold value for a subsequent stage. Further, outputs from two or more preceding stages may be combined together to provide a threshold value for a current stage. Although four gain/filter stages are shown in these drawings, fewer or more gain/filter stages may exist. Output signal 514D from demodulator 510D will be used to provide an output value for the circuit.

[0033] The architecture of FIG. 5 A can also be modified, so that a single mixer is used before the gain/filter stages and the differential signals provided by the single mixer pass through a single set of gain/filter stages. Such an embodiment is depicted in FIG. 5C. In this drawing, amplifier 506 receives differential signals from antenna 512 and provides an amplified signal to mixer 502, which mixes the differential input signals with 0° and 180° differential signals from local oscillator 520, then provides differential signals to gain/filter stages 508A, 508B, 508C, 508D. The differential output signals from each gain/filter stage are sent to a successive gain/filter stage, and also to a respective quadrature phase shifter 522A, 522B, 522C, and 522D. Quadrature phase shifters 522A, 522B, 522C, and 522D are passive asynchronous quadrature

phase shifters, an embodiment of which is depicted in FIG. 6 and described hereinbelow. Quadrature phase shifter 600 operates as a low loss, all-pass filter (i.e., no filtering is performed) but a phase shift is performed so that the differential inputs IN+ and IN. become quadrature outputs OUTi+, OUTi., OUTQ+, and OUTQ-. These quadrature outputs 524A, 524B, 524C, 524D are provided as input to demodulators 510A, 510B, 5 IOC, 510D. As described hereinabove, an output signal from each demodulator may be provided to a successive demodulator in the string of demodulators, i.e., demodulator 510A may provide output signal 514A to demodulator 510B; demodulator 51 OB may provide output signal 514B to demodulator 5 IOC; and demodulator 5 IOC may provide output signal 514C to demodulator 510D. The output signal from one stage may be used as a threshold value for a subsequent stage or combined with the output of additional stages to provide a threshold value for a current stage. Again, output signal 514D from demodulator 510D will be used to provide an output value for the circuit. In the circuits of FIGS. 5A-5C, all elements are DC compatible, i.e. the elements are coupled to each other without needing capacitors between the elements.

[0034] FIG. 6 depicts passive asynchronous quadrature phase shifter 600 according to an embodiment. Quadrature phase shifter 600 forms a closed loop that includes resistors Rl, R2, R3, R4 and capacitors CI, C2, C3, C4. Capacitor CI is coupled between resistors Rl and R2; capacitor C2 is coupled between resistors R2 and R3; capacitor C3 is coupled between resistors R3 and R4; and capacitor C4 is coupled between resistors R4 and Rl . Differential signal IN+ and IN. are input to quadrature phase shifter 600, with input signal IN+ being coupled between Rl and C I and also being coupled between R4 and C4. Similarly, input signal IN. is coupled between R2 and C2 and also between R3 and C3. Quadrature output signals OUTi+ and OUTi. are taken between Rl and C4 and between R3 and C2 respectively; similarly, OUTQ+, OUTQ. are taken between R2 and CI and between R4 and C3 respectively. As described hereinabove, a copy of quadrature phase shifter 600 can be provided between each gain/filter stage and a respective demodulator.

[0035] In one embodiment of FIG. 5C, quadrature phase shifters 522 are mixer-based, rather than based on the cyclic arrangement of resistors and capacitors of FIG. 6. Although not specifically shown, such an embodiment, would use an arrangement similar to that shown by mixers 502A, 502B and local oscillators 520A, 520B. When used as quadrature phase shifters, this mixer/oscillator approach would be configured to up-convert each of differential phase

signals 516A, 516B, 516C, 516D using quadrature phase clocks to generate four different phases OUTi+, OUTi-, OUTQ+, OUTQ.. This arrangement leads to a lower area requirement than the embodiment of FIG. 6 in that mixers use transistors, which are small, rather than the larger resistors and capacitors required by the .resistor/capacitor embodiment.

[0036] FIG. 7A discloses another block diagram of a low power wake-up receiver 700A. In this drawing, mixers 702 receive differential radio frequency input signal 701. Mixer 702A mixes 0° and 180° signals from local oscillator 704A with RF signal 701, and mixer 702B mixes 90° and 270° signals from local oscillator 704B with RF signal 701. The output from mixers 702A, 702B is provided to transimpedance amplifiers 706A and 706B, where amplification of the signal is performed, and then to polyphase filter 708 where filtering is performed, such as to remove blocker frequencies. Polyphase filter 708 then provides the amplified, filtered quadrature signals to both demodulator 710 and peak detector 712. Although not specifically shown in this drawing, the output from peak detector 712 will also be provided to demodulator 710. This embodiment shares similarities with FIG. 5 A, so it is not extensively described herein. This embodiment is primarily provided to illustrate how this architecture can be used to support further improvement in the speed of peak detection, as shown in FIG. 7B, which further improves the speed of peak detection by providing an additional phase shift of 45 degrees. This is done by using a broadband frequency divider that provides all the 45° phases to the mixers. Using the additional 45° granularity, this embodiment can obtain an eight times speed improvement of the peak detection time over single-phase operation, thereby greatly improving the energy consumption of the system.

[0037] Wake-up receiver 700B in FIG. 7B is essentially a doubled version of the circuit of FIG. 7A, with mixer 702 A mixing 0° and 180° signals from local oscillator 704A with RF signal 701; mixer 702B mixing 90° and 270° signals from local oscillator 704B with RF signal 701; mixer 702C mixing 45° and 225° signals from local oscillator 704C with RF signal 701; and mixer 702D mixing 135° and 315° signals from local oscillator 704D with RF signal 701. The output from mixers 702A, 702B, 702C and 702D is provided respectively to transimpedance amplifiers 706A, 706B, 706C and 706D and then to polyphase filters 708A and 708B respectively. Polyphase filter 708A then provides the quadrature signals to both demodulator 710A and peak detector 712, while polyphase phase-filter 708B provides the quadrature signals to demodulator 710B and peak detector 712. Additional granularity in phase can also be added using further similar hardware blocks providing simultaneous signal processing.

[0038] Accordingly, example embodiments include wireless receiver architectures that use quadrature phases of the input signal to provide faster signal recognition while using low power and can also use threshold values that are generated internally, rather than being provided by an outside source. Advantages of the described architectures may include one or more of the following:

• A mixer-first approach leads to signal processing at low power, reducing power consumption;

• The input impedance of the receiver can be very large without a noise penalty, leading to low power consumption;

• The front-end gain of the receiver architecture leads to a reduction of noise from subsequent stages; thereby large resistance values can be used to lower area and power consumption;

• After down-conversion, N phases at an intermediate frequency (IF) are applied to the peak-detector, leading to a N times speedup in detection time;

• More precision and faster detection can be obtained by making the gain of these stages accurate; at subthreshold region gm a 1/RVHSR leads to a very precise realization of gain; The peak detector can also be linearized to improve linear detection range;

• More detection speed (lower latency) can be achieved by adding similar hardware in parallel paths with more phase granularity

• The entire system is fully on-chip, and can use self-calibration for robust operation;

• The example embodiments are fully integrated, with low power and low area;

• Largely process insensitive by using multiple sections of polyphase signal, and using signal dependent thresholds for comparison.

[0039] In this description, reference to an element in the singular does not mean "one and only one" unless explicitly so stated, but rather "one or more."

[0040] Modifications are possible in the described embodiments, and other embodiments are possible, within the scope of the claims.