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1. (WO2019032047) A CIRCULARLY POLARIZED ANTENNA FOR RADIO FREQUENCY ENERGY HARVESTING
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A Circularly Polarized Antenna for Radio Frequency Energy Harvesting

Field

The present solution relates to a circularly polarized antenna for Radio Frequency energy harvesting, for example, a dual band circularly polarized antenna for Radio Frequency (RF) energy harvesting.

Background

Energy harvesting, also known as energy scavenging is a method that converts surrounding electromagnetic waves into electrical energy. Recently, there is an increase in the literature in Radio Frequency (RF) energy harvesting application in areas such as low power wireless sensors, radio frequency identification (RFID) tags, and biotelemetry. Periodic battery replacements for large amount of sensor nodes are unrealistic and expensive. Hence, scavenging of ambient RF energy for wireless sensor network (WSN) is gaining popularity at least in the literature written on it.

Specifically, slotted circular microstrip disk antennas are described in, "Comparative Study of Antenna Designs for RF Energy Harvesting", Hindawi Publishing Corporation, International Journal of Antennas and Propagation, Publication Date: 2013, Authors: Sika Shrestha, Sun-Kuk Noh, and Dong- You Choi, Vol. 2013, Article ID 385260, and in, "Compact Circularly Polarized Rectenna With Unbalanced Circular Slots", IEEE Transactions on Antennas and Propagation, Vol. 56, No. 3, March 2008, Authors: Tzong-Chee Yo, Chien-Ming Lee, Chen-Ming Hsu, and Ching-Hsing Luo. A microstrip patch antenna with irregular slots is proposed in, "A new compact size microstrip patch antenna with irregular slots for handheld GPS application" University of Iraq, Eng. & Technology, Vol.26, No. 10, Published in 2008, Author: Jawad K. Ali. However, these 3 literatures describe antennas working in a single band and the gain achieved by each of the proposed antennas is less than 5.0 dBic.

Summary

According to an example of the present disclosure, there are provided a method and apparatus as claimed in the independent claims. Some optional features are defined in the dependent claims.

Brief Description of the Drawings

Examples of the present disclosure will be better understood and readily apparent to one skilled in the art from the following written description, by way of example only and in conjunction with the drawings, in which:

Figure 1 A illustrates a cross-sectional view of an antenna based on an example of the present disclosure.

Figure 1 B illustrates a top view of a stacked slotted-circular-patch (SCP) in Figure 1 A Figure 1 C illustrates a top view of a tapered-slit-octagon patch (TSOP) in Figure 1 A. Figure 1 D illustrates a top view of a microstrip feed line in Figure 1 A.

Figure 1 E illustrates a photograph of a top view of a prototype antenna according to an example of the present disclosure.

Figure 2A illustrates measured and simulated return loss of the antenna of Figure 1 A. Figure 2B illustrates measured and simulated axial-ratio at the boresight of the antenna of Figure 1 A.

Figure 2C illustrates measured and simulated gain at the boresight of the antenna of Figure 1 A.

Figure 2D illustrates a normalized radiation pattern for plane xz at 910 MHz of the antenna of Figure 1 A.

Figure 2E illustrates a normalized radiation pattern for plane yz at 910 MHz of the antenna of Figure 1 A.

Figure 2F illustrates a normalized radiation pattern for plane xz at 91 1 MHz of the antenna of Figure 1 A.

Figure 2G illustrates a normalized radiation pattern for plane yz at 91 1 MHz of the antenna of Figure 1 A.

Figure 2H illustrates a normalized radiation pattern for plane xz at 2.43 GHz of the antenna of Figure 1 A.

Figure 2I illustrates a normalized radiation pattern for plane xz at 2.41 GHz of the antenna of Figure 1 A.

Figure 3A illustrates a dual band rectifier with a voltage doubler circuit configuration according to an example of the present disclosure.

Figure 3B illustrates a dual-band composite right/left handed (CRLH) transmission line (TL) impedance matching circuit according to an example of the present disclosure. Figure 3C shows a specific configuration of the composite right/left handed (CRLH) transmission line (TL) impedance matching circuit of Figure 3B.

Figure 4 shows a graph regarding reflection coefficient of the rectifier of Figure 3A.

Figure 5 shows a photograph of a top view of a prototype dual band circularly polarized (CP) rectenna according to an example of the present disclosure.

Figure 6A shows a graph regarding simulated and measured RF-DC conversion efficiency for various values of load resistance at f = 900 MHz for a dual band rectenna according to an example of the present disclosure.

Figure 6B shows a graph regarding simulated and measured RF-DC conversion efficiency for various values of load resistance at f = 2.45 GHz for a dual band rectenna according to an example of the present disclosure.

Detailed Description

Typically, a Radio Frequency (RF) harvester will harvest ambient RF energy and store the energy in micro-batteries to power the WSN. With advances and popularity of wireless communication devices, large amount of abundant RF energy from surrounding sources are scattered in our environment. Using an appropriate antenna, these electromagnetic waves can be converted into electrical energy. There is present a challenge in that linearly polarized antenna receives only noise signals when the antenna is not aligned with the existing electromagnetic waves. Also, ambient electromagnetic waves may exist in all sorts of orientation and polarization. Therefore, an example of the present disclosure proposes a circularly polarized (CP) antenna for RF energy harvesting application. A CP antenna is insensitive to multi-path effects and is able to harness RF energy regardless of the orientation of the device comprising the CP antenna. In the example, a dual band CP rectenna is proposed to harvest RF energy at GSM (Global System for Mobile communication) and WiFi (Wireless Fidelity) bands over a wide coverage efficiently. In the present disclosure, GSM (Global System for Mobile communication) band and WiFi (Wireless Fidelity) band refer to GSM frequency band and WiFi frequency band respectively. The CP rectenna has been shown can receive most of the ambient RF energy/waves (linearly polarized, circularly polarized, elliptically polarized, slanted polarized waves) from the surrounding/environment.

When harvesting RF energy at the UHF band, there is a need for size reduction to let the RF energy harvesting apparatus or system remain compact. Symmetrical-slit and asymmetrical-slit methods are examples of the present disclosure proposed to achieve apparatus or system size reduction to handle energy harvesting from CP radiation or RF signals. It has been observed that a truncated corners CP microstrip antenna does not offer size reduction to a radiator for energy harvesting while

asymmelric-slit microstrip antenna can be used for size reduction to handle energy harvesting from CP radiation or RF signals.

An example of the present disclosure provides a compact dual-band circularly polarized (CP) antenna for energy harvesting. Specifically, the antenna is a multilayered stacked dual-band compact CP microstrip antenna with wide coverage along with a metamaterial based rectifier. Specifically, in the present example, the metamaterial based rectifier is a compact dual band rectifier that comprises a dual band matching network realized using a Composite Right/Left Hand (CRLH) transmission line based circuit, which constitutes the metamaterial. The antenna is proposed for RF energy harvesting applications operating at the GSM and Wi-Fi bands.

The antenna of the present example further comprises a tapered-slit-octagon patch (TSOP) with a proximity coupled-feed and stacked slotted-circular-patch (SCP) fed by a metallic-via (i.e. electrical connection) connecting to a microstrip feed line. The TSOP is constructed by embedding eight tapered-slits arranged symmetrically on the octagon patch along octagonal axes from a center of the patch to reduce the patch size and to generate two orthogonal modes with equal magnitude for CP radiation or RF signals. Each of the eight tapered-slits has different length, and the tapered slits have a gradual length reduction among them. In one example, a tapered slit's length is reduced by about 6.25% of a length of an adjacent tapered slit. In one example, the minimum length reduction percentage between two tapered slits (not necessary adjacent to each other) of the TSOP is 0%. The maximum length reduction percentage between two tapered slits (not necessary adjacent to each other) of the TSOP is 44%. In another example, one tapered slit may have no length reduction or 0% length reduction, the maximum tapered slit length reduction for one tapered slit is 43.75% length reduction, and there is a gradual length reduction percentage of 6.25% between every two adjacent tapered slits. The gradual length reduction percentage between every two adjacent tapered slits may range from 0% to 6.35%.

In the present example, the SCP is constructed by two asymmetric-circular slots with, for example, a ratio of 1 :6 embedded diagonally on the circular patch to generate two orthogonal modes with equal magnitude for CP radiation or RF signals. A dual-band composite right/left handed (CRLH) based rectifier operating at 900MHz and 2.45 GHz may be assembled using a dual-band matching circuit and a voltage doubler circuit. The dual-band matching circuit may be designed using a compact CRLH transmission line (TL) that is able to produce two different phase shifts at two different frequencies. In the present example, the antenna can achieve measured gain of more than 5.2 dBic across the band of 908.0 MHz● 922.0 MHz and gain of more than 6.14 dBic across the band of 2.35 GHz● 2.50 GHz with peak gains of 5.41 dBic at 91 8.0 MHz and 7.94 dBic at 2.485 GHz. An overall antenna volume of the present example is 0.36● οx0.36●οx0.026● 0 (●0 is free space wavelength at 900 MHz). The rectifier size is 0.1 8●οx0.075●οx0.0002●ο at 900 MHz and measured RF-DC conversion efficiency is 43% at 900 MHz and 39% at 2.45 GHz.

In summary, the present example provides a slit-patch radiator i.e. the TSOP for wide coverage receipt of RF signals at GSM band and a slotted-patch radiator i.e. the SCP is for receiving RF signals at WiFi band. An advantage of the present example is that a wide-coverage is achieved at GSM band due to the slits in patch radiator i.e. the TSOP.

Figure 1 A illustrates a cross-sectional view of the antenna 100 based on the aforementioned example.

In the present example, the antenna 1 00 may be configured with an overall antenna size (volume) of 120.0 mm (length) x 1 20.0 mm (width) x 8.6 mm (height) . The antenna 1 00 comprises three substrates, a top substrate 1 01 , a middle substrate 1 03 and a bottom substrate 1 05. The three substrates 1 01 , 1 03 and 1 05 in the present example are materials or dielectric materials suitable for antenna design. In the present example, the three substrates 1 01 , 1 03 and 1 05 may be known as dielectric substrates. A SCP 1 02 (R0 = 1 7.6 mm) is disposed on top of the top substrate 1 01 . The top substrate 1 01 has a height of = 1 .6 mm, dielectric constant of εr = 3.4, and tanδ = 0.002. A radiating TSOP 1 04 (with La = 80.1 mm and Sw = 4.0 mm; more details on these parameters will be provided later) is disposed on top of the middle substrate 1 03 (which can be regarded as an upper dielectric layer). Specifically, the TSOP 1 04 is disposed between the top substrate 101 and the middle substrate 1 03. The middle substrate 1 03 has a height of h2 = 5.4 mm, dielectric constant of εr =3.4, and tan<5>" = 0.002. A microstrip feed line 108 (of 50-● ) with a width of 3.67 mm is printed top on the bottom substrate 1 05. Specifically, the microstrip feed line 1 08 is disposed between the middle substrate 1 03 and the bottom substrate 1 05. The bottom substrate 105 has a height of h1 = 1 .6 mm. dielectric constant of εr = 3.4, and tanδ = 0.002. The bottom substrate 1 05 is disposed above a ground plane 1 1 0. The SCP 1 02 is fed by a via 1 06 connecting the SCP 1 02 to the microstrip feed line 108. The TSOP 104 comprises a Via hole 1 1 2 for the via 106 to go through to connect the SCP 1 02 to the microstrip feed line 108 as the TSOP 1 04 is located between the SCP 1 02 and the microstrip feed line 1 08. The via 106 should not contact the TSOP 1 04. The design of the present example is such that the SCP 1 02 and TSOP 1 04 are being proximity coupled-fed by the microstrip feed line 1 08. The SCP 1 02 and the microstrip feed line 1 08 are connected but the TSOP 1 04 is not connected.

In Figure 1 A, the ground plane 1 1 0 covers entirely a bottom of the bottom substrate 1 05. In top view, the SCP 102 is centrally located at a center of the antenna 100. For example, Figure 5 shows a clear top view that the SCP 102 may be located at such central location of the antenna 100. In the present example, the TSOP 104 is configured to span across almost close to an entire length and width of the antenna 1 00. In another example, a via hole 1 12 need not be provided and the via 1 06 may be routed around the TSOP 104 to connect the SCP 102 and the microstrip feed line 1 08 instead of going through the via hole 1 1 2.

Figure 1 B shows a top view of the SCP 1 02 of Figure 1 A. A vertical y axis and a horizontal x axis are shown in Figure 1 B to provide reference for the configuration of the SCP 102. The SCP 1 02 is circular in shape with a diameter of 2Ro. The SCP 1 02 comprises two asymmetric-circular slots 1 1 4 and 1 16 centered at c1 and c2 respectively. The circular slot 1 14 has a diameter of 2r1 (r1 being the radius). The circular slot 1 1 6 has a diameter of 2r2 (r2 being the radius). The circular slots 1 14 and 1 1 6 are disposed with, for example, a ratio of 1 :6, and disposed diagonally along a line 1 1 1 of symmetry of the SCP 1 02. The ratio 1 :6 means, for instance, that if the circular slot 1 14 has a diameter of about 1 mm, the circular slot 1 1 6 would have a diameter of about 6mm. The circular slots 1 14 and 1 16 are disposed on the SCP 1 02 such that two orthogonal modes with equal magnitude for CP radiation or RF signals would be generated. The line 1 1 1 forms an acute angle● with the y axis.

Figure 1 C shows a top view of Figure 1 A revealing the TSOP 1 04 located below the top substrate 1 01 . The TSOP 104 is substantially octagonal in shape with a side to side length of La and a side length of about Lc. Eight tapered-slits (hereinafter collectively or individually referenced using reference numeral 1 1 3) are each arranged symmetrically on the TSOP 1 04 at each of the eight respective octagonal axes stemming from a center T of the TSOP 104. Each of the eight octagonal axes intersects the center T of the TSOP 1 04 and a vertex of the TSOP 1 04. Figure 1 C shows one octagonal axis 1 16 of the eight octagonal axes. Each of the eight tapered-slits 1 1 3 is symmetrically disposed at each of the 8 vertices of the TSOP 1 04. A slit width Sw, which is a parameter for

characterizing slit size, is a distance between a vertex of the TSOP 1 04 to a point of the TSOP 1 04 that is closest to the vertex. For example, Sw is shown in Figure 1 C to be a distance between a vertex 1 20 to a point 1 1 8 of the TSOP 104 that is closest to the vertex 1 20.

Furthermore, each of the eight tapered-slits 1 1 3 is of different length from one another. The length of each of the eight tapered-slits in the present example is taken to be a distance from the respective vertex of the TSOP 1 04 to a point of the TSOP 1 04 that is along the octagonal axis intersecting the vertex and closest to the center 7 of the TSOP 1 04. For example, length of a slit at the vertex 1 20 is a distance from the vertex 120 to a point P8 of the TSOP 1 04 that is along the octagonal axis 1 1 6 and closest to the center 7 of the TSOP 104. The tapered slits 1 1 3 are configured to have a gradual length reduction among them. For instance, in Figure 1 C, each of the eight tapered-slits has an apex point closest to the center 7 of the TSOP 104. The apex points are shown to be adjacent to one another in consecutive order of P1 to P8 in Figure 1 C. In the present example, the slit with P8 has the shortest length and the slit with P1 has the longest length. In one example, the tapered slit length of one tapered slit is a gradual reduction of about 6.25% of the length of an adjacent tapered slit. The via hole 1 1 2 of Figure 1 is shown to be present between the slit with apex point P4 and the slit with apex point P3.

Figure 1 D shows a top view of Figure 1 A revealing the microstrip feed line 1 08 located below the bottom substrate 105. In Figure 1 D, length, L, and width, W, of the antenna 100 are shown. Length, S, and width, ml, of the microstrip feed line 1 08 are also marked out in Figure 1 D. The microstrip feed line 108 is an elongate line located about midway of the width, W, of the antenna 1 00 and a first end of the line 1 08 ends on one side of the antenna 100. The via 1 06 is located close to a second end of the line 1 08 opposite to the first end.

With reference to Figures 1 A to 1 D, the antenna design dimensions of the parameters marked out in Figures 1 A to 1 D and detailed information of the proposed antenna design of the present example are given in Tables I, II, and III below. The Figure in which the parameters can be found in or are featured in are indicated in the Tables. Equations for x and y refers to respective x and y coordinates taken with respect to the respective x and y axes found in Figures 1 B and 1 C respectively.

Table I - Slit and slot ositions



The values of the above-mentioned dimensions and parameters are optimized for one specific example of a dual band (GSM and WiFi) CP antenna, it should be appreciated that in other examples, the operating frequency can be tuned/changed within the GSM and WiFi bands, and correspondingly, there can be different variations in the values of the antenna design dimensions and parameters. Figure 1 E illustrates a photograph of a top view of a prototype antenna according to an example of the present disclosure. Specifically, the prototype antenna is a dual-band CP antenna. If the prototype antenna of Figure 1 E has the configuration of the example described with reference to Figures 1 A to 1 D, the SCP 102 of the antenna 100 of Figure 1 A, and c1 and C2, which are the centers of the two asymmetric-circular slots 1 14 and 1 16, would appear as marked out in Figure 1 E.

Figure 2A shows a graphical comparison 201 of measured and simulated return loss of the dual-band CP antenna 100 with the design as described with reference to Figures 1 A to 1 D. Specifically, Figure 2A shows a graphical comparison of measured and simulated return loss of the dual-band CP antenna 100 against frequency in GHz. In particular, the measured 10-dB return loss bandwidth of 46.0 MHz (889 MHz●935 MHz) and 165.0 MHz (2.360 GHz●2.525 GHz), and the simulated 10-dB return loss bandwidth 30.0 MHz (888 MHz●918 MHz) and 156.0 MHz (2.349 GHz●2.505 GHz). The graph shows a good agreement between simulated and measured return loss of the antenna 100.

Figure 2B shows a graphical comparison 202 of measured and simulated axial-ratio at the bores ight of the dual-band CP antenna 100 with the design as described with reference to Figures 1 A to 1 D. Specifically, Figure 2B shows the measured and simulated axial ratio of dual-band CP antenna 100 against frequency in GHz. The Simulated 3-d B AR bandwidths are 12.0 MHz (893 MHz●905 MHz) and 20.0 MHz (2.41 GHz●2.43 GHz). The measured axial ratio data's in dual-band appears to follow the simulated curves.

Figure 2C shows a graphical comparison 203 of measured and simulated gain at the boresight of the dual-band CP antenna 100 with the design as described with reference to Figures 1 A to 1 D. Specifically, Figure 2C shows the measured and simulated gain of the dual-band CP antenna against frequency in GHz. It is observed that the antenna 100 exhibits a simulated gain of more than 5.5 dBic across the band of 900.0 MHz● 910.0 MHz, and a gain of more than 6.8 dBic across the band of 2.360 GHz● 2.550 GHz with peak gains of 5.57 dBic at 906.0 MHz and 7.87 dBic at 2.490 GHz. Furthermore, the antenna 100 exhibits the measured gain of more than 5.2 dBic across the band of 908.0 MHz● 922.0 MHz and gain of more than 6.14 dBic across the band of 2.350 GHz● 2.500 GHz with peak gains of 5.41 dBic at 918.0 MHz and 7.94 dBic at 2.485 GHz. Both measured and simulated gain data illustrate good agreement.

Figure 2D to 2I shows measured normalized radiation patterns of the dual-band CP antenna 100 with the design as described with reference to Figures 1 A to 1 D at GSM-band and Wi-Fi band. Specifically, Figure 2D shows a normalized radiation pattern 204 for planes xz at 910 MHz. with a 3-dB axial ratio beam width that is around 180° in the plane xz at GSM-band. Figure 2E shows a normalized radiation pattern 205 for plane yz at 910 MHz, with a 3-dB axial ratio beam width that is around 180° in the plane yz at GSM-band. Figure 2F shows a normalized radiation pattern 206 for plane xz at 91 1 MHz, with a 3-dB axial ratio beam width that is around 180° in the plane xz at GSM-band. Figure 2G shows a normalized radiation pattern 207 for planes yz at 91 1 MHz, with a 3-dB axial ratio beam width that is around 180° in the plane yz at GSM-band. Figure 2H shows normalized radiation pattern 208 at 2.43 band in a plane xz and good AR (less than 3 dB) can be seen at round boresight at Wi-Fi band. Figure 2l shows a normalized radiation pattern 209 at 2.43 band a plane xz and good AR (less than 3 dB) can be seen at round boresight at Wi-Fi band.

Figures 3A to 3C illustrate an example of a dual-band rectifier structure and design that is proposed to work with the antenna 100 of Figures 1 A to 1 D. In the present example, the dual band rectifier 300 is designed to operate at 900 MHz and 2.45 GHz. It comprises a dual-band CRLH based impedance matching circuit 302 and a voltage

doubler circuit 304. The dual-band CRLH based impedance matching circuit 302 is configured with a 2-unit cell CRLH TL constructed using a combination of microstrip line of electrical length and lumped components. The CRLH TL is of compact design and is able to produce two different phase shifts at two different frequencies i.e. 900 MHz and

2.45 GHz.

Specifically, Figure 3A shows an example of the configuration of a voltage doubler circuit 304. The voltage doubler circuit 304 is connected to a power source comprising of the impedance matching circuit 302 and the antenna 100. In the present example, the power source is designed to operate at 900 MHz and 2.45 GHz. In the present example, the voltage doubler circuit 304 comprises a first capacitor C 301 of 100pF. A first plate of the first capacitor C 301 is connected to the power source. A second plate of the first capacitor C 301 is connected to a first diode 303 (for example, SC7630) and a second diode 305 (for example, SC7630). The cathode end of the first diode 303 is connected to the second plate of the first capacitor C 301 . The anode end of the first diode 303 is connected to ground. The anode end of the second diode 305 is connected to the second plate of the first capacitor C 301 and the cathode end of first diode 303. A first plate of the second capacitor C 307 of 100pF is connected to ground and a second end of the second capacitor C 307 is connected to the second diode 305. The second end of the second capacitor C 307 is connected to the cathode end of the second diode 305 and a resistor R1 309 of 500 ohm. Output voltage Vout 310 is across the resistor R1 309. A first end of the resistor R1 309 is connected to the second capacitor C 307 and a second end of the resistor R1 309 is connected to ground.

Figure 3B shows an example of the configuration of the impedance matching circuit 302. The impedance matching circuit 302 comprises a first microstrip line 306 connected to the microstrip feed line 108 of the antenna 100. In one example, the first microstrip line 306 can be the microstrip feed line 108. A first plate of the third capacitor 2CI 31 1 is connected to the first microstrip line 306 and a second plate of the third capacitor 2CI 31 1 is connected to a first end of a first inductor LI 313. A second end of the first inductor LI 313 is connected to ground. The first end of the first inductor LI 313 and the second end of the third capacitor 2CI 31 1 are connected to a first plate of the fourth capacitor CI 315. A second plate of the fourth capacitor CI 315 is connected to a first end of a second inductor LI 317. A second end of the second inductor LI 317 is connected to ground. The first end of the second inductor LI 317 and the second end of the fourth capacitor CI 315 are connected to a first plate of a fifth capacitor 2CI 319. A second piate of the fifth capacitor 2CI 319 is connected to a second microstrip line 308. The second microstrip line 308 is connected to the first plate of the first capacitor C 301 of the voltage doubler circuit 304. The capacitance of each of the third capacitor 2CI 31 1 and fifth capacitor 2CI 319 is twice of that of the capacitance of the fourth capacitor CI 315. Each of the first microstrip line 306 and the second microstrip line 308 has no phase difference and their power angle● R are the same. Each of the first inductor LI 313 and the second inductor LI 317 has the same inductance value. Figure 3C shows an example comprising proposed values for the components of the impedance matching circuit 302 of Figure 3B. For example, the first microstrip line 306 and the second microstrip line 308 each has a length of 27.1264 mm and a width of 0.667 mm. The capacitance of the fourth capacitor CI 315 is 19.097 pF. The capacitance of each of the third capacitor 2CI 31 1 and fifth capacitor 2CI 319 is twice of that of the capacitance of the fourth capacitor CI 31 5. Each of the first inductor LI 313 and the second inductor LI 317 has the same inductance value of 127.86 nH.

Figure 4 illustrates a graph 400 showing reflection coefficient S1 1 of the dual-band rectifier 300 described with reference to Figures 3A to 3C for an input power of 0 dBm. The reflection coefficient S1 1 is shown to be less than 20dB at both the desired frequency bands (i.e. 900 MHZ and 2.45 GHz), which the dual-rectifier 300 is designed to operate in.

Figure 5 shows a top view of a prototype of a rectifier with the antenna, which can be known as a circularly polarized rectenna 500. A rectenna refers to a rectifying antenna, which is a special type of receiving antenna that is used for converting electromagnetic energy into direct current (DC) electricity. If the circularly polarized rectenna 500 is configured according to the example described with reference to the antenna 100 of Figures 1 A to 1 D and dual band rectifier 300 of Figures 3A to 3C, the antenna 100, the SCP 102 of the antenna 100, and the dual band rectifier 300 would appear as marked up in Figure 5.

An example of the present disclosure is a rectenna comprising the the antenna 100 of Figures 1 A to 1 D and dual band rectifier 300 of Figures 3A to 3C like the prototype of Figure 5. Figure 6A shows a graph 600 indicating simulated and measured RF-DC conversion efficiency for various values of load resistance i.e. RI = 500 ohms, 1000 ohms or 1500 ohms at GSM-band i.e. f = 900 MHZ for such dual-band rectenna. The RF-DC conversion efficiency shown in Figure 6A achieves more than 40% efficiency. Figure 6B shows a graph 602 indicating simulated and measured RF-DC

conversion efficiency for various values of load resistance i.e. R, = 500 ohms, 1000 ohms or 1500 ohms at Wi-Fi band i.e. f = 2.45 GHz for such dual-band rectenna. The RF-DC conversion efficiency shown in Figure 6B achieves more than 40% efficiency.

The proposed examples of the circularly polarized (CP) antenna described in the present disclosure or the more specifically described dual-band rectenna based on the various examples described herein may be implemented in low powered wireless sensors for harvesting ambient RF energy. The antennas of the proposed examples can be scalable for energy harvesting in other industrial, scientific and medical (ISM) radio bands like 400 MHz, 5.5 GHz and even 60 GHz. The solution might further find application in TV White space, GPS, UHF RFID and Chipless RFID in addition to just RF harvesting.

Although specific parameter values have been provided for examples of a circularly polarized (CP) antenna (e.g. 100 of Figures 1 A to 1 D) and a dual band rectifier (e.g. 300 of Figures 3A to 3C), it is appreciated that other examples are not limited to these specific parameter values and suitable values that would enable the circularly polarized (CP) antenna and dual band rectifier to work are also possible.

The proposed antenna in the examples of the present disclosure is low profile and miniature (i.e. compact). It has a dual-band high gain design (for example, greater than 5 dBic), wherein size and dimensions would be big issues. Therefore, the prospect and chances of the commercialization of such proposed antenna is high.

Examples of the present disclosure may have the following features.

A circularly polarized (CP) antenna (e.g. 100 of Figure 1 A) for Radio Frequency energy harvesting comprising: a top substrate (e.g. 101 of Figure 1 A); a bottom substrate (e.g. 105 of Figure 1 A) disposed on a ground plane (e.g. 1 10 of Figure 1 A); a middle substrate (e.g. 103 of Figure 1 A) disposed between the top substrate and the bottom substrate; a slotted patch (e.g. 102 of Figure 1 A) comprising more than one slots, the slotted patch being disposed on the top substrate; a slitted patch (e.g. 104 of Figure 1 A) comprising a plurality of slits (e.g. 1 13 of Figure 1 C), the slitted patch being disposed between the top substrate and the middle substrate; a microstrip feed line (e.g. 108 of Figure 1 A) disposed between the middle substrate and the bottom substrate; and a via (e.g. 106 of Figure 1 A) connecting the slotted patch and the microstrip feed line to enable the slotted patch to be fed by the microstrip feed line, wherein the more than one slots and the slits are arranged so as to generate more than one orthogonal modes with equal magnitude for receiving Radio Frequency (RF) signals.

The slitted patch (e.g. 1 04 of Figure 1 A) may comprise a via hole (e.g. 1 12 of Figure 1 A) for the via (e.g. 1 06 of Figure 1 A) to pass through the slitted patch.

The top substrate (e.g. 1 01 of Figure 1 A) and bottom substrate (e.g. 105 of Figure 1 A) may have same thickness and the middle substrate (e.g. 1 03 of Figure 1 A) may have a thickness thicker than the thickness of the top substrate and bottom substrate.

The slotted patch (e.g. 1 02 of Figure 1 A) may be a siotted-circular-patch comprising two circular slots (e.g. 1 14 and 1 1 6 of Figure 1 B).

The slotted patch may comprise two circular slots (e.g. 1 14 and 1 16 of Figure 1 B) with a size ratio of 1 :6.

Each of the plurality of slits (e.g. 1 1 3 of Figure 1 C) may be tapered and the plurality of slits may be symmetrically arranged in the slitted patch (e.g. 1 04 of Figure 1 A).

The slitted patch (e.g. 1 04 of Figure 1 A) may be octagonal in shape and each of the plurality of slits (e.g. 1 1 3 of Figure 1 C) may be disposed at a location of a vertex (e.g. 1 20 of Figure 1 C) of the slitted patch.

Each of the plurality of slits (e.g. 1 13 of Figure 1 C) may be cut from an edge of the slitted patch (e.g. 104 of Figure 1 A) towards a center (e.g. T of Figure 1 C) of the slitted patch and the plurality of slits may be cut at different length from one another.

The plurality of slits (e.g. 1 1 3 of Figure 1 C) may have a difference in length of 0% to 6.35% between adjacent slits.

The received Radio Frequency (RF) signals may be in GSM (Global System for Mobile communication) band and/or WiFi (Wireless Fidelity) band.

A dual-band circularly polarized (CP) antenna (e.g. 1 00 of Figure 1 A) for Radio Frequency energy harvesting comprising: a top dielectric substrate (e.g. 1 01 of Figure 1 A) ; a bottom dielectric substrate (e.g. 1 05 of Figure 1 A) disposed on a ground plane (e.g. 1 10 of Figure 1 A) ; a middle dielectric substrate (e.g. 103 of Figure 1 A) disposed between the top dielectric substrate and the bottom dielectric substrate; a slotted-circular-patch (e.g. 102 of Figure 1 A) comprising two circular slots (e.g. 1 14 and 1 1 6 of Figure 1 B), the slotted-circular-patch being disposed on the top dielectric substrate; a tapered-slit-octagon patch (e.g. 1 04 of Figure 1 A) comprising eight tapered slits (e.g. 1 1 3 of Figure 1 C) of different lengths symmetrically arranged in the tapered-slit-octagon patch and each of the eight tapered slits is disposed at a location of a vertex (e.g. 1 20 of Figure 1 C) of the tapered-slit-octagon patch, the tapered-slit-octagon patch being

disposed between the top dielectric substrate and the middle dielectric substrate; a microstrip feed line (e.g. 1 08 of Figure 1 A) disposed between the middle dielectric substrate and the bottom dielectric substrate; and a via (e.g. 1 06 of Figure 1 A) connecting the siotted-circular-patch and the microstrip feed line to enable the slotted-circular-patch to be fed by the microstrip feed line, wherein the slitted patch comprises a via hole (e.g. 112 of Figure 1 A) for the via to pass through the slitted patch, wherein the two circular slots and the eight tapered slits are arranged so as to generate two orthogonal modes with equal magnitude for receiving Radio Frequency (RF) signals in GSM (Global System for Mobile communication) band and WiFi (Wireless Fidelity) band , wherein the top dielectric substrate and bottom dielectric substrate have same thickness and the middle dielectric substrate has a thickness thicker than the thickness of the top dielectric substrate and bottom dielectric substrate.

A circularly polarized (CP) rectenna (e.g. 500 of Figure 5) comprising: the aforementioned circularly polarized (CP) or dual-band circularly polarized (CP) antenna; a composite right/left handed (CRLH) transmission line (TL) impedance matching circuit (e.g. 304 of Figure 3A) ; and a voltage doubler circuit (e.g. 302 of Figure 3B or 3C).

The CRLH TL impedance matching circuit (e.g. 304 of Figure 3A) may be configured to produce two different phase shifts at two different frequencies.

The CRLH TL impedance matching circuit (e.g. 304 of Figure 3A) may comprise metamaterial based CRLH TL.

Throughout this specification and claims which follow, unless the context requires otherwise, the word "comprise", and variations such as "comprises" or "comprising", will be understood to imply the inclusion of a stated integer or group of integers or steps but not the exclusion of any other integer or group of integers.

While the invention has been described in the present disclosure in connection with a number of embodiments and implementations, the invention is not so limited but covers various obvious modifications and equivalent arrangements, which fall within the purview of the appended claims. Although features of the invention are expressed in certain combinations among the claims, it is contemplated that these features can be arranged in any combination and order.